Physical Methods of Speed-Independent Module Design english
Physical Methods of Speed-Independent Module Design english
Physical Methods of Speed-Independent Module Design
Any method of logic circuit design is based on using formal models of gates and wires. The simplest model of a gate
is determined by only two "parameters": (a) Boolean function is to be calculated, (b) fixed propagation delay. The simplest model of a wire is an
ideal medium with zero resistance and consequently, with zero delay. Such simple models allow circuit design procedures which are a sequence of
elementary steps easily realized by a computer.
When logic circuits designed by using the simplest models expose unreliable operation as in the case of gate delay
variations, designers introduce less convenient but more realistic models with arbitrary but finite delay. Using more complicated models may produce logic
circuits that are called speed-independent .
In speed-independent circuits transition duration can be arbitrary. So a centralized clock cannot be used. Instead
special circuitry to detect output validity is applied. Besides, additional interface circuitry is needed to communicate with the environment in a
handshaking manner. A speed-independent circuit can be seen as a module consisting of combinational logic (CL) proper, CL output validity detector
(OVD) and interface circuitry (Fig.1). To enable OVD to distinguish valid output data from invalid ones, the redundant coding scheme was proposed .
The main idea of the scheme is to enumerate all possible input and output data, both valid and invalid. The OVD must be provided with appropriate information
on data validity. To realize the idea of redundant coding some constraints on CL design are imposed :
(i) CL must be free of delay hazards, i.e. CL output data word must not be dependent on the relative
delay of signal paths through CL.
(ii) In changing between input states, any intermediate or transient states that are passed through must
not be mapped by CL onto valid output states.
When these constraints were formulated, the circuit designers realised that not every Boolean description could be
implemented in a speed-independent style. Other approaches to speed-independent module design were needed.
SIM design as a science has two branches: logical and physical. For a long time physical branch was overshadowed in spite of its competitiveness. The main properties of physical approach to
SIM design are:
(a) Arbitrary coding scheme.
(b) Conventional procedure of operational unit design.
(c) Races of signals in SIM do not affect on its proper operation.
In this paper we propose an approach based on the physical nature of transitions in CL. We believe that each
transition is actually a transfer of energy which can be naturally detected by physical methods.
From the viewpoint of a radio engineer CL behaves like a radio transmitter. It emits radio frequencies in the 108-1010Hz
band modulated by signals of 106-108Hz. Obviously, the carrier wave is produced by gate switchings during transitions in CL. The
modulating wave is produced by control schemes (OVD and interface circuitry) that detect transition completion and inform the environment about the
readiness of CL. OVD is a kind of radio receiver that extracts the modulation envelope and enhances the received signal. The main properties that OVD circuit
must expose from a radio engineer's point of view are selectivity and high gain. Since the useful signals can propagate through non-conducting medium, OVD
circuits can be coupled with CL indirectly.
Advances in semiconductor technology gave birth to two methods of transition detecting based on two kinds of the
information carrying signal, namely electromagnetic radiation and current consumption. Frequency of the signal produced by switching logic gates is
determined by gate delay.
For instance, CMOS network of 1-ns gates produces 1-GHz signal, ECL array of
100-ps gates gives 10-GHz radiation. Logic circuits consisting of 10-ps gates will emit infra-red radiation. That signal could be easily detected by photosensitive devices.
Let us have a closer look at the structure of speed-independent modules (SIM) as presented in Fig.1. All input
data are processed in CL, all output data are obtained from CL, too. So, CL is the only unit in SIM which is involved in proper data processing. The result of
that processing is specified by Boolean functions. Algorithms for calculating the Boolean function are realised by the internal structure of CL. Generally,
its structure is series-parallel as well as algorithm implemented.
When n-bit data word is put into the CL, n or more signal propagation paths (SPPs) can be activated
concurrently. So, one can say that the calculation of a Boolean function by CL is of parallel nature. On the other hand, each SPP is a gate chain which
processes data in a serial manner. So, calculation in CL is also of sequential nature.
The OVD circuit is intended for detecting transient and steady "states" of CL. If any SPP in CL is still
"active", CL is in transient state, otherwise it is in steady state. Each gate switching results in both logical and electromagnetic effects on its
surrounding medium. The logical effects of switching has been heavily investigated; we consider physical one.
To provide speed-independence of the module the OVD and interface circuitry must also work in a speed-independent
mode. This means that any arbitrary but finite transistor or wire delay cannot impair proper operation of OVD and interface circuitry.
The interface circuitry is a mediator between OVD and environment of SIM. It implements any kind of signalling
convention, commonly a two- or four-cycle one  based on request Req and acknowledgement Ack signal using. The interface circuitry receives the
output validity (OV) signal from the OVD circuit, a Req signal from the environment and transmits an Ack signal to the environment (Fig.1).
Consider an algorithm of operation for interface circuitry realizing speed-independent four-cycle signalling
convention (FCSC). In accordance with FCSC the control signals must go in the following sequence: Req+®OV-®Ack+®Req-®Ack- where "+"
corresponds to rising the signal and "-" corresponds to falling the signal. All signals are assumed to adhere to positive logic. Initially the
signals Req and Ack are low, the signal OV is high. If the environment state changes, the Req signal rises and transient state of
CL occurs (OV-). Upon completion of the transitions in CL, signal OV rises and the interface circuitry generates the Ack
signal rising. After that the environment produces a falling Req signal and then the interface circuitry transmits the falling Ack signal to the
environment. All the signals have to be reset into the initial state.
To develop the interface circuitry a circuit designer must take into account that any OVD circuit has finite
(non-zero) turn-on delay ton. This means that OVD cannot respond on transitions of short duration t tr< ton
An example of interface circuitry is shown in Fig.2. It contains a flip-flop, a NOR-gate, an asymmetrical delay and
an inverter as an output stage .
The asymmetrical delay is intended for delaying Req rising signal for t+ period where t+ > ton
. Delaying Req falling signal noted t- is to be as short as
possible. Note that speed-independent operation of interface circuitry is vulnerable to delay t+ variation. If t+ becomes less than ton
, proper operation of SIM can not be guaranteed. Otherwise, if t+ is much more than ton
, performance of SIM will be significantly reduced. To provide exact accordance of t+ and ton
a circuit emulator can be used.
Such an emulator is either an exact copy of OVD or its functional copy, i.e. resistive-capacitive model of OVD's
critical path. In the chip the emulator must be placed next to active OVD circuit in order to ensure identical conditions of fabrication and operation.
In this example we use a simplified asymmetrical delay implemented as an asymmetrical CMOS inverter chain (Fig.3). Contrary to
the common inverter an asymmetrical one has non-equal rise and fall times of output signal.
A time diagram for interface circuitry is presented in Fig.4 for two cases: (a) ttr < ton
and (b) ttr ³ ton. In case (a) the signal sequence Req+®Ack+ is formed for (t++tNOR)
period where tNOR is a NOR-gate delay. In case (b) the above sequence is formed for (ttr +toff+tNOR)
duration where toff is a turn-off delay of OVD circuit. When the SIM returns to the initial steady state, the signal sequence Req-®Ack- is formed for (t-+tNOR)
After considering the SIM in operation it is obvious that the main problems of the module design are in the area of CL
and OVD interaction. This includes (a) kind of signal used as a carrier of information about CL output validity, and (b) method of OVD circuit design.
4. Current consumption detection
Using current consumption of CMOS CL for output validity detection was proposed in 1990 . Contrary to the method of
EMR detection this one is based on introducing direct coupling of source and receiver. While CL is in steady state it consumes current of about 10-9-10-8A
which does not allow OVD switching. The interface circuitry gets information on CL output validity and in turn informs the environment about CL readiness to
input data processing. When an input data arrives CL changes its state to "transient", current consumption increases to 10-4-10-2A,
which switches the OVD, thus informing the interface circuitry about output invalidity. The latter lets the environment know about CL business.
After the computations in the CL are finished, the current consumption decreases down to the steady
state value, and the OVD sends a signal of output validity.
4.1 Information carrying signal
Current consumption by CMOS CL contains useful information on CL state. CMOS CL is a network of CMOS gates, so
the current consumed by CL is a superposition of currents consumed by CMOS gates included in the CL. Each CMOS
gate contains PMOS transistor and NMOS transistor networks (Fig.5). While a gate is in a steady state either the PMOS or the NMOS network is in a
conducting mode. When a gate switches the non-conducting transistor network becomes conducting. There is usually a short period in switching time when both
networks are in a conducting mode.
Generally, current consumed by a CMOS gate includes three components [9,10]:
(a) leakage current Ilk passing between power supply and ground due to finite
resistance of non-conducting transistor network;
(b) short-circuit current Isc flowing while both networks are in a conducting mode;
(c) load capacitance CL charge current ILC flowing
while a CMOS gate is switching from low to high output voltage via conducting PMOS network and CL .
SPICE simulation has shown  that amplitude of current consumed by a typical CMOS inverter depends on CL
and is limited by the non-zero resistance of the conducting PMOS network (Fig.7). The integral of consumed current is proportional to CL
. When a gate switches from high to low output voltage, the component ILC
is negative by direction and negligible by value (Fig.7b). It is evident, the switchings from high to low output voltage occur at the expense of energy
accumulated in CL during the previous switching from low to high output voltage. The component Isc does not depend on direction in which a gate
The component ILC equals to ILC = CL×Vdd ×f where Vdd is a power supply
voltage, f is a gate switching frequency. Veendrick has investigated the component Isc dependencies on CL and
rise-fall time of input potential signal . He showed that if both input and output signal have the same rise-fall time, the component Isc
cannot be more than 20 percent of summary current consumption . However, when the output signal rise-fall time is less than input one, the component Isc
can be of the same order of magnitude as ILC. In that case it must be taken into account. As to the component Ilk, it
entirely depends on CMOS process parameters and for state of the art CMOS devices Ilk is about 10-15 -10-12 A.
So, the analysis of CMOS gate current consumption allows us to conclude that in transient state a CMOS gate consumes
a current IS= Ilk+Isc+ILC
and in steady state it consumes only Ilk<< IS . The difference
between two states from the viewpoint of current consumption is several orders of magnitude. So, CMOS gate output validity detection is possible, both in
principle and in practice.
In Section 2 we presented series-parallel model of computations in CL. We showed that in every moment during switching
current consumed by CL is a superposition of the currents consumed on the activated signal propagation paths (SPPs). Now, considering CL implemented by
CMOS devices we should note that while logical signal propagates through SPP the neighbouring gates switch in opposite directions. That is why a curve of
current consumed by a ten inverter chain (Fig.8) looks like a combination of crests and troughs. Nevertheless, in the very lowest point of the curve the
current consumed by CL in a transient state remains several orders more than in a steady state.
4.2 OVD implementation
The proposed OVD circuit, shown in Fig.9, is a threshold circuit translating an
analog current signal I into a logical signal OV.
The OVD circuit contains a current-to-voltage converter (CVC) consisting of the resistor R1 and
the diode D1. The OVD also contains a comparator implemented by the MOS transistors M1-M7 and resistors R2,,,R3
. CMOS CL consumes the current I and introduces a capacitance Cin
. The capacitance Cout represents the load caused by the interface circuitry. A low potential output signal of OVD corresponds to CL
output validity. A high potential output signal corresponds to CL output invalidity. So, OVD generates OV signal in negative logic manner.
The transfer characteristics of CVC is determined by a system of three equations:
where I is an input current of CVC, DV is a voltage drop on
the CVC circuit, Ir is a current flowing through the resistor R1, Id is a current passing through the
diode D1, I0 is a leakage current of the diode, rb is a bulk resistance of the diode. Here stands for kT/q where k is Boltzmann's constant, T is absolute temperature, q is charge
of an electron.
Equations (1)-(3) determine the functional connection F between input current I and voltage drop DV: . Graphic solution of the system is shown in Fig.10.
CVC parameters to be calculated are R1 and rb. Initial data for calculating R1 are
the threshold voltage drop DVth and corresponding threshold input
current Ith . Value Ith is determined by minimal current consumed by CMOS CL in transient state. Initial data for
calculating rb are maximal voltage drop DVmax and corresponding
maximal input current Imax. Value Imax is determined by the maximal number of gates in CL switching simultaneously and
their load capacitances.
The comparator chosen is the CMOS ECL receiver proposed by Chappell et al.. The circuit includes a single
differential amplifier stage with built-in compensation for parameter variations, followed by a CMOS inverter. The comparator has 100-mV worst-case
sensitivity in 1-mm technology. Detailed static and dynamic analysis of the comparator circuit was given in .
The comparator compares input voltage signal Vin with reference voltage Vref. If Vin <Vref
the comparator output signal equals to logical zero which means that CL outputs are valid. Otherwise, Vin >Vref,
the comparator output signal equals to logical "one" which means that the outputs are invalid.
As it follows from the OVD circuit configuration,
where Vdd is a voltage of power supply.
Equations (4) and (5) allow us to calculate the threshold voltage drop DV of the CVC circuit:
since , so
If 0<DV<500mV then the diode D1 of CVC operates in the very small current region Id
» 0 and Id <<Ir. So the component Id in the Equation (1) can be neglected and I»Ir =DV/R1 .
For practical values of the threshold input current of the OVD circuit is reversely proportional
to the resistance of R1 : . Substituting Equation (6) yields
As to choosing value of rb it must be done with regard to maximal voltage drop DVmax .
If DV>750mV, the diode D1 is in active mode and while rb <<R1
the condition Ir <<Id is true. So, in the large current region I»Id and Equation (2)
determines an almost linear dependence between I and DV. For instance, if the
maximal voltage drop DVmax =900mV and maximal input current Imax=2mA,
then in accordance with the Equation (2) rb »100W. Typical element values
for the OVD circuit with DVth =400mV are given in Table 1.
The turn-on ton and turn-off toff delays of the OVD circuit depend on the OVD
itself and the CMOS CL as well. (Switching the OVD output from low to high voltage is called "turning-on" and reverse switching is called
Consider a piece of CMOS CL and its interaction with OVD circuit (Fig.11). The piece is an SPP including N
logic gates. Each gate is shown symbolically as a connection of PMOS and NMOS networks. All the capacitances
affecting ton and toff can be brought down to three components:
(i) CLi is the load capacitance of the i-th gate;
(ii) Cpsi is the power supply bus capacitance associated with the i-th gate;
(iii) Cin is the input capacitance of the OVD circuit.
Let pi is a probability of the i-th gate being in the state of high output potential. In this
state the capacitance CLi is connected with power supply bus through the low channel resistance of turned-on transistors in PMOS network of
the i-th gate. Then equivalent capacitance Ceq connected to the OVD circuit input equals
where N is a number of gates in the considered SPP. Here the resistance of conducting PMOS network is assumed to be
Equation (7) is also true for CL including several SPPs. In that case summing must be
carried out for all the gates belonging to CL.
Simulation shows that ton and toff are proportional to the OVD time constant t=R1Ceq.
It was also obtained that when N>20, the component under the sign of summation in Equation (7) can be much larger than the component Cin.
Due to voltage drop DV the effective power supply voltage is reduced
and CL performance is decreased by about 35 percent .
In order to make SIM operating faster special attention must be paid to reducing the capacitance introduced by CL.
4.3 Speed-independent address bus
The simplest case of CL is a scheme degenerated into a set of wires called
a multi-bit bus. Let us develop the OVD circuit for such a CL.
Multi-bit bus consists of several lines. Each line can be considered as a medium
for signal propagating from one end of the chip to another. Delay of signal propagation through a
line depends on several factors:
(a) output impedance and symmetry of driver circuit;
(b) initial state of the line: if driver is symmetrical, line switching from high to low voltage
lasts shorter than reverse switching;
(c) electrical properties of the line as a signal propagation medium (resistance of conducting layer and
capacitances between the line and other wires next to it);
(d) length of the line;
(e) input impedance and sensitivity of receiving circuit.
Since different lines of the bus operate in different conditions (a)-(e), signal propagation delays are
different, too. From the standpoint of environment the bus behaves like any other more complicated CL.
Asynchronous RAM designers use a bus transition detector since 1980s [13-15]. Such a detector is usually based on double-rail address coding and two series
connected transistors for each address bit . One of the transistors receives the true address signal
and the other receives the complementary address signal of the particular address bit. For any steady state condition one of the transistors will be
turned on and one will be turned off. There will be a finite rise and fall time during a transition of the
address bit. There is a short time during which both transistors are conducting. The establishment of the conductive path provides the detection
address transition. In the first asynchronous RAMs the output signal of the transition detector is used for bit
line precharging and for enabling/disabling sense amplifiers and peripheral circuitry.
Self-timed RAM announced in 1983  used transition detectors not for address transition only but also for detecting read/write completion
and address/bit line precharge completion as well.
The CMOS transition detector was invented in 1986 . This circuit is also based on double-rail coding and uses a pair
of series-connected NMOS transistors (Fig.12). The scheme for n-bit bus control contains n line
transition detectors (LTDs) and n AND-gates. Outputs of AND-gates are united in node M forming
wired OR. The output inverter serves as a pulse shaper. Capacitors C1 and C2 are intended to prolong rise time of the LTD output
signal (true and complementary). This is necessary for reliable detection.
The main drawback of the circuit is speed dependence. One can see that if true and complementary address bit
signal have different propagation delays, the conducting path via NMOS transistors will never be formed.
Using the OVD circuit proposed in Section 4.2 as LTD we can avoid this drawback.
Note that address transmission through the address bus is unidirectional. So
to detect completion of bus transition it is enough to recognize the bus state at the destination end. For this
purpose we modify CL to consist of n lines. The modification means introducing n LTDs, each actually a CMOS inverter chain. Each chain
contains two inverters loaded with a capacitance (Fig.13). Input of each LTD is connected with corresponding line of the bus at the destination end. Power
supply pads of all LTDs are connected to the current input of the same OVD circuit.
The parameters of the input current signal for the OVD circuit are varied
(i) value of capacitances C1 and C2 ;
(ii) dimensions of MOS transistors M1 -M4 .
Since all transitions in CL are of the same duration and can be lengthened to be outlast the OVD turning-on time, we simplify the interface circuitry by disallowing the asymmetrical delay.
Due to short duration of normal transition in this CL we must take into
account the integral nature of the sensitivity of the OVD circuit. OVD sensitivity depends on both amplitude and width of input current pulse.
Simulated operation region of the OVD circuit for current pulses shorter than 30ns is shown in Fig.14. It is obvious that in this case the threshold of the
OVD circuit must be determined by threshold charge Qth value. The OVD input charge Q equals to where I is OVD input current, t is a moment of
time when transition occurs, w is a width of input current pulse. Turning-on condition for the OVD circuit is Q=Qth.
When the LTD circuit shown in Fig.13 is used, the charge value Q is determined by either C1 or C2.
Namely, if the line goes from low to high voltage, Q=dV×C2. If the line goes in
the reverse direction then where dV is charging/discharging
voltage, approximately equal to the effective power supply voltage: dV»Vdd -DV. Here Vdd
is OVD power supply voltage and DV is CVC voltage drop.
The OVD circuit with typical parameters (See Table 1) has a threshold charge value Qth =4.0×10-12 C. When
C1 =C2 =CL , the minimal value of CL providing OVD capacity for operation is about 1.0×10-12 F.
Influence of transistors M1 -M4 dimensions on LTD
delay d is determined by approximation :
where ~ is a sign of proportionality, Gn and Gp are
the conductances of NMOS and PMOS transistors respectively (CL =C1 =C2.)
Since and where W and L are width and length of transistor channels of the corresponding conduction type, the LTD delay d
is proportional to .
It has been obtained that for , , CL=1.0pF and Vdd-DV=5.0V the LTD delay d=7.6ns.
When LTD works jointly with the OVD in the speed-independent bus, the real value of the LTD delay will increase
by 30-40 percent due to OVD's R1 effect on the effective power supply voltage.
To determine the appropriate value of R1 in the OVD circuit we must know threshold input current Ith
corresponding to threshold voltage drop DVth recommended to be equal
Average input current Iav in transient state of one line is determined by the expression Iav =CL×v where
v is the average rate of increase in the output signal for an inverter included in LTD. For typical
values v=1.0×109 Volts per second and CL =1.0pF, Iav
=1.0mA. Accepting Ith =0.4mA and Imax=2.0mA we obtain R1=1kW and rb=100W.
Simulation has shown that in this case OVD turning-on delay can be approximated by an empirical expression:
where n is the address bus bit capacity. Total delay of recognizing address transition ttot =d×g+ton
where g is a coefficient of the LTD delay increase due to reducing power supply voltage. As we showed above g»1.35. It can be seen
that if n=32, ttot=21.6ns.
4.4 Speed-independent adder
The circuit we use in this Section as a CL was a touch-stone for many speed-independent circuit designers for about
four decades. We mean a ripple carry adder (RCA) which is actually a chain of one-bit full adders (Fig.14).
Each full adder calculates two Boolean functions: sum si=aiÅbiÅci and output carry ci+1=aibi+bici+aici where ai, bi are summands, ci is input
carry and Å stands for XOR operation.
In 1955 Gilchrist et al. proposed speed-independent RCA with carry completion signal . In 1960s that circuit
was carefully analyzed and improved [19-21]. In 1980 Seitz used RCA for illustrating his concept of equipotential region and his approach to self-timed
system design .
Now we use RCA as a CL for illustrating our approach to SIM design.
As it was shown in Section 4.2 the turn-on and turn-off delays of the OVD circuit are proportional to the equivalent
capacitance Ceq associated with OVD circuit input. Capacitance Ceq depends linearly on a number of gates N
in CMOS CL. To speed up a SIM it is necessary to reduce a number N. This can be reached by structural decomposition CMOS CL into subcircuits CL1, CL2,
etc. Each subcircuit CLi is connected to its own detecting circuit OVDi or directly to the power supply if this subcircuit transition does not
affect the transition duration in CL as a whole. Each detecting circuit OVDi generates its own OV signal which is combined with other OVDs' output
signals via a multi-input OR (NOR) element. The output signal of that element serves as OV signal of the CMOS CL.
Multi-bit RCA computation time is determined by length of maximal activated carry chain. A lot of papers were
devoted to analysis of carry generation and carry propagation in RCA [19-21], many of them contained their own methods for estimation or calculation of
average maximal activated carry chain. We do not intend to add another one.
Let us have a look inside RCA. As it was mentioned above RCA consists of one-bit full adders and each full adder consists of two parts: forming
sum si part and forming carry ci+1 part (Fig.16).
In multi-bit RCA all forming sum parts do not interact with each other and do not
affect on transition duration in RCA. Each forming carry ci+1 part receives ci
signal from preceding forming carry part and sends ci+1 signal
to consequent one.
To decompose RCA we use three heuristic tricks:
(i) All forming sum parts we connect directly to power supply.
(ii) We divide each forming carry part into three subcircuits denoted in Fig.16 by numbers 1,2
All subcircuits 1 we connect directly to power supply because they do not contain input ci and so do not contain carry propagation
(iii) All subcircuits 2 we connect to OVD1 and all subcircuits 3 we connect to OVD2. Outputs
of OVD1 and OVD2 are connected to two-input NOR-gate forming RCA OV signal in positive logic manner
OVD1 and OVD2 input currents I1 and I2 curves for 6-bit RCA and longest transition duration
are shown in Fig.18.
Accepting DVth1,2=400mV we calculated the
OVD circuits parameters. It was obtained R11=5kW, Ith1=0.08mA,
OVD1 and OVD2 delay dependencies on a number of bits in RCA are shown in Fig.19.
4.5 Comparison of SIMs with synchronous counterparts
Transition duration in CL is a random variable. Probability of transition with duration D
is determined by implemented Boolean function and distribution of input logical combinations. Domain of possible values for
variable D occupies the interval [0;Dmax]. Here Dmax is a length of critical
path in CL.
Let is a mathematical expectation of transition
duration in CL where Di is a length of i-th SPP in CL, pi is a
probability of i-th path being the longest activated SPP.
When CL works in the synchronous mode, the cycle duration Ts is chosen with regard to maximal
transition duration Dmax. Certain margin must be added to Dmax
to provide reliable operation of CL in the case of CL parameter variations: Ts
=k×Dmax where k is a
In SIM cycle duration is a random variable with expectation Tsi = g×Dme+toff+tif
where g is a coefficient of CL delay increasing due to reducing power supply voltage, toff
is turn-off delay of the OVD circuit, tif is an interface circuitry delay.
We determine efficiency E for speed-independent mode of CL operation as relative increase of SIM performance
in comparison to its synchronous counterpart:.
Generally, speed-independent mode is more efficient than synchronous one if Ts >Tsi
or, in other words, .
In the case of RCA where tc is a delay of
carry forming part, n is a number of full adders in RCA.
It has been shown  that in n-bit RCA Dme» tc×log2(5n/4).
Then, in the case of speed-independent operation Tsi=g×tc×log2(5n/4)+toff+tif.
We have obtained dependencies of Ts , Tsi
on a number of bits in RCA that are shown in Fig.20. As it can be seen, speed-independent
operation of RCA is more efficient while n>8.
I would like to thank Igor Shagurin and Vlad Tsylyov of the Moscow Physical Engineering Institute for
helpful discussions of this work. I am also grateful to Chris Jesshope of University of Surrey and Mark Josephs of Oxford University who kindly provided
the latest material on their research in the area of delay-insensitive circuit design.
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SPICE2G.6: MOSFET model parameters
ZERO-BIAS THRESHOLD VOLTAGE
BULK THRESHOLD PARAMETER
DRAIN OHMIC RESISTANCE
SOURCE OHMIC RESISTANCE
ZERO-BIAS B-D JUNCTION
ZERO-BIAS B-S JUNCTION
BULK JUNCTION SATURATION
BULK JUNCTION POTENTIAL
GATE-SOURCE OVERLAP CAPACI-
TANCE PER METER CHANNEL WIDTH
GATE-DRAIN OVERLAP CAPACI-
TANCE PER METER CHANNEL WIDTH
GATE-BULK OVERLAP CAPACITANCE
PER METER CHANNEL LENGTH
DRAIN AND SOURCE DIFFUSION
ZERO-BIAS BULK JUNCTION BOTTOM
CAPACITANCE PER SQ METER OF
BULK JUNCTION BOTTOM GRADING
ZERO-BIAS BULK JUNCTION SIDE-
WALL CAPACITANCE PER METER OF